Method and system for pre-equalization in a single weight (SW) single channel (SC) multiple-input multiple-output (MIMO) system

ABSTRACT

In wireless systems, a method and system for pre-equalization in a single weight (SW) single channel (SC) multiple-input multiple-output (MIMO) system are provided. A first receive antenna and at least one additional receive antenna may receive a plurality of SC communication signals transmitted from at least two transmit antennas. Estimates of the propagation channels between transmit and receive antennas may be performed concurrently and may be determined from baseband combined channel estimates. Channel weights may be determined to modify the signals received by the additional receive antennas. Pre-equalization weight parameters may be determined to modify subsequent signals transmitted from the transmit antennas. The pre-equalization weight parameters may be based on the propagation channel estimates and may be determined by LMS, RLS, DMI, or by minimizing a cost function. Closed loop transmit diversity may also be supported.

CROSS-REFERENCE TO RELATED APPLICATIONS/INCORPORATION BY REFERENCE

This patent application is a continuation of application Ser. No.11/172,702 filed on Jun. 30, 2005. This patent application makesreference to, claims priority to and claims benefit from U.S.Provisional Patent Application Ser. No. 60/616,733 filed on Oct. 6,2004.

This application makes reference to:

-   U.S. patent application Ser. No. 11/173,870 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/174,303 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,502 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,871 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,964 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,252 filed Jun. 30, 2005,    issued as U.S. Pat. No. 7,471,694 on Dec. 30, 2008;-   U.S. patent application Ser. No. 11/174,252 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/172,756 filed Jun. 30, 2005,    issued as U.S. Pat. No. 7,552,562 on Apr. 21, 2009;-   U.S. patent application Ser. No. 11/173,305 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/172,759 filed Jun. 30, 2005,    issued as U.S. Pat. No. 7,483,675 on Jan. 27, 2009;-   U.S. patent application Ser. No. 11/173,689 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,304 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,129 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/172,779 filed Jun. 30, 2005,    issued as U.S. Pat. No. 7,586,886 on Sep. 8, 2009;-   U.S. patent application Ser. No. 11/173,727 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,726 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/172,781 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/174,067 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,854 filed Jun. 30, 2005;-   U.S. patent application Ser. No. 11/173,911 filed Jun. 30, 2005; and-   U.S. patent application Ser. No. 11/174,403 filed Jun. 30, 2005,    issued as U.S. Pat. No. 7,505,539 on Mar. 17, 2009.

Each of the above referenced applications is hereby incorporated hereinby reference in its entirety.

FIELD OF THE INVENTION

Certain embodiments of the invention relate to the processing ofwireless communication signals. More specifically, certain embodimentsof the invention relate to a method and system for pre-equalization in asingle weight (SW) single channel (SC) multiple-input multiple-output(MIMO) system.

BACKGROUND OF THE INVENTION

In most current wireless communication systems, nodes in a network maybe configured to operate based on a single transmit and a single receiveantenna. However, for many of current wireless systems, the use ofmultiple transmit and/or receive antennas may result in an improvedoverall system performance. These multi-antenna configurations, alsoknown as smart antenna techniques, may be utilized to reduce thenegative effects that multipath and/or signal interference may have onsignal reception. Existing systems and/or systems which are beingcurrently deployed, for example, CDMA-based systems, TDMA-based systems,WLAN systems, and OFDM-based systems such as IEEE 802.11a/g/n, maybenefit from configurations based on multiple transmit and/or receiveantennas. It is anticipated that smart antenna techniques may beincreasingly utilized both in connection with the deployment of basestation infrastructure and mobile subscriber units in cellular systemsto address the increasing capacity demands being placed on thosesystems. These demands arise, in part, from a shift underway fromcurrent voice-based services to next-generation wireless multimediaservices that provide voice, video, and data communication.

The utilization of multiple transmit and/or receive antennas is designedto introduce a diversity gain and/or an array gain and to suppressinterference generated within the signal reception process. Suchdiversity gains improve system performance by increasing receivedsignal-to-noise ratio, by providing more robustness against signalinterference, and/or by permitting greater frequency reuse for highercapacity. In communication systems that incorporate multi-antennareceivers, a set of M receive antennas may be utilized to null theeffect of (M-1) interferers, for example. Accordingly, N signals may besimulataneously transmitted in the same bandwidth using N transmitantennas, with the transmitted signal then being separated into Nrespective signals by way of a set of N antennas deployed at thereceiver. Systems that utilize multiple transmit and multiple receiveantenna may be referred to as multiple-input multiple-output (MIMO)systems. One attractive aspect of multi-antenna systems, in particularMIMO systems, is the significant increase in system capacity that may beachieved by utilizing these transmission configurations. For a fixedoverall transmitted power, the capacity offered by a MIMO configurationmay scale with the increased signal-to-noise ratio (SNR).

However, the widespread deployment of multi-antenna systems in wirelesscommunications, particularly in wireless handset devices, has beenlimited by the increased cost that results from increased size,complexity, and power consumption. Providing a separate RF chain foreach transmit and receive antenna is a direct factor that increases thecost of multi-antenna systems. Each RF chain generally comprises a lownoise amplifier (LNA), a filter, a downconverter, and ananalog-to-digital converter (A/D). In certain existing single-antennawireless receivers, the single required RF chain may account for over30% of the receiver's total cost. It is therefore apparent that as thenumber of transmit and receive antennas increases, the systemcomplexity, power consumption, and overall cost may increase.

In the case of a single RF chain with multiple antennas, there is theneed to determine or estimate separate propagation channels. A simplemethod may comprise switching to a first receive antenna utilizing, forexample, an RF switch, and estimate a first propagation channel. Afterestimating the first propagation channel, another receive antenna may beselected and its corresponding propagation channel may be estimated. Inthis regard, this process may be repeated until all the channels havebeen estimated. However, switching between receive antennas may disruptthe receiver's modem and may lower throughput. This approach may requireadditional hardware and may also result in propagation channel estimatesat different time intervals. Any mechanisms that may be utilized tocompensate for the presence of multiple time-varying propagationchannels may also present added complexity and cost to the design andoperation of MIMO systems.

A single weight approach may work best for a single path, that is, forflat fading channels, because a single weight may not combine all pathsarriving at different delays optimally. To optimally combine eachmultipath at receiving antennas may require multiple weights atdifferent delays. For example, the same number of weights as multipathsarriving at different delays may be required, which may be more like acomplete channel equalization approach. On the other hand, utilizing asingle weight may have an average combining effect on multiple paths,with sub-optimal performance. A single weight may not be selected sothat an optimized combination of multiple paths may be achieved at thereceiving antennas. For example, for a Rayleigh flat fading channel, asingle weight solution may result in about a 6 dB gain, while for thechannels with many Rayleigh faded paths the gain may be reduced to about2 dB.

Moreover, multi-path propagation in band-limited time dispersivechannels may cause inter-symbol interference (ISI), which has beenrecognized as a major obstacle in achieving increased digitaltransmission rates with the required accuracy. ISI may occur when thetransmitted pulses are smeared out so that pulses that correspond todifferent symbols are not discernable or separable. Meanwhile, datareceived from a desired user may be disturbed by other transmitters, dueto imperfections in the multiple access scheme, giving rise tointer-carrier interference (ICI). For a reliable digital transmissionsystem, it is necessary to reduce the effects of ISI and ICI.

Further limitations and disadvantages of conventional and traditionalapproaches will become apparent to one of skill in the art, throughcomparison of such systems with some aspects of the present invention asset forth in the remainder of the present application with reference tothe drawings.

BRIEF SUMMARY OF THE INVENTION

A method and/or system for pre-equalization in a single weight (SW)single channel (SC) multiple-input multiple-output (MIMO) system,substantially as shown in and/or described in connection with at leastone of the figures, as set forth more completely in the claims.

These and other advantages, aspects and novel features of the presentinvention, as well as details of an illustrated embodiment thereof, willbe more fully understood from the following description and drawings.

BRIEF DESCRIPTION OF SEVERAL VIEWS OF THE DRAWINGS

FIG. 1A is a block diagram of an exemplary two-transmit (2-Tx) andtwo-receive (2-Rx) antennas wireless communication, in accordance withan embodiment of the invention.

FIG. 1B is a block diagram of an exemplary two-transmit (2-Tx) andtwo-receive (2-Rx) antennas wireless communication withpre-equalization, in accordance with an embodiment of the invention.

FIG. 1C is a block diagram of an exemplary two-transmit (2-Tx) andtwo-receive (2-Rx) antennas wireless communication system that supportsWCDMA/HSPDA, in accordance with an embodiment of the invention.

FIG. 1D is a block diagram of an exemplary two-transmit (2-Tx) andtwo-receive (2-Rx) antennas wireless communication system withpre-equalization that supports WCDMA/HSPDA, in accordance with anembodiment of the invention.

FIG. 1E is a block diagram of an exemplary two-transmit (2-Tx) andmultiple-receive (M-Rx) antennas wireless communication system withpre-equalization that supports WCDMA/HSPDA, in accordance with anembodiment of the invention.

FIG. 2A is a flow diagram illustrating exemplary steps for channelestimation in a 2-Tx and M-Rx antennas wireless communication system, inaccordance with an embodiment of the invention.

FIG. 2B illustrates an exemplary periodic phase rotation for an in-phase(I) signal received in one of the additional receive antennas, inaccordance with an embodiment of the invention.

FIG. 3A is a block diagram of an exemplary single weight basebandgenerator (SWBBG) that may be utilized in a 2-Tx and 2-Rx antennassystem, in accordance with an embodiment of the invention.

FIG. 3B is a block diagram of an exemplary single weight basebandgenerator (SWBBG) that may be utilized for channel pre-equalization in a2-Tx and M-Rx antennas system, in accordance with an embodiment of theinvention.

FIG. 3C is a block diagram of an exemplary RF phase and amplitudecontroller, in accordance with an embodiment of the invention.

FIG. 4 is a flow diagram illustrating exemplary steps in the operationof the single weight baseband generator (SWBBG) that may be utilized ina 2-Tx and M-Rx antennas system, in accordance with an embodiment of theinvention.

FIG. 5 is a flow diagram illustrating exemplary steps for determiningchannel weights in additional receive antennas utilizing signal-to-noiseratio (SNR) or signal-to-interference-and-noise ratio (SINR), inaccordance with an embodiment of the invention.

DETAILED DESCRIPTION OF THE INVENTION

Certain embodiments of the invention may be found in a method and systemfor pre-equalization in a single weight (SW) single channel (SC)multiple-input multiple-output (MIMO) system. A first receive antennaand at least one additional receive antenna may receive a plurality ofSC communication signals transmitted from at least two transmitantennas. Estimates of the propagation channels between transmit andreceive antennas may be performed concurrently and may be determinedfrom baseband combined channel estimates. Channel weights may bedetermined to modify the signals received by the additional receiveantennas. Pre-equalization weight parameters may be determined to modifysubsequent signals transmitted from the transmit antennas. Thepre-equalization weight parameters may be based on the propagationchannel estimates and may be determined by LMS, RLS, DMI, or byminimizing a cost function. Closed loop transmit diversity may also besupported. The various embodiments of the invention may provide a goodcompromise between implementation complexity and performance gains toreduce the effects of, for example, inter-symbol interference (ISI)and/or inter-carrier interference (ICI) in MIMO systems.

Most communication channels suffer from multipath fading. To addressmultipath fading, different equalizer techniques may be used. Generally,equalization algorithms may be implemented at the receiver side of acommunication link. However when the equalizer weight solution isavailable at the transmitter, then pre-equalizer techniques may be used.The method of equalization may be the same for pre-equalization at thetransmitter and post-equalization at the receiver when the optimalweights become the inverse conjugate of the channel, for example. Theweights may therefore be applied at either the transmitter duringpre-equalization or at the receiver during post-equalization. Theseweights may be optimal when there is no interference present in thesystem. When any interference sources are present in the system, theoptimum weights for the pre-equalization and post-equalization may bedifferent. One of the benefits of using pre-equalizer techniques lies inthe simplification of the receiver architecture that results from movingthe complexity of the equalization operation to the transmitter.However, pre-equalizer techniques may be related to the feedback of thechannel estimates. The delay may cause some lag between the receivedsymbols and the corresponding transmitted symbols. Pre-equalizationweights may be calculated for vector and matrix channels and applied tothe transmitted symbols accordingly.

An approach that supports channel pre-equalization at the transmittermay be utilized to improve upon the use of a single weight solution whenmultipath signals are received by multiple receive antennas. The purposeof pre-equalization is to make signals appear at the receiver as asingle path, that is, a flat fading channel. In the case of flat fadingchannels the two receive antenna SW solution may yield maximum gain thatmay approach 6 dB gain in flat fading Rayleigh channels, for example. Aproblem with pre-equalization with multiple receive antenna systems isthat the pre-equalization may not pre-equalize the channels optimallyfor all multiple receiving antennas. An averaging effect may occur,which pre-equalizes the multipath channel at the multiple receivingantennas partially.

FIG. 1A is a block diagram of an exemplary two-transmit (2-Tx) andtwo-receive (2-Rx) antennas wireless communication, in accordance withan embodiment of the invention. Referring to FIG. 1A, there is shown awireless communication system 100 that may comprise a first transmitantenna (Tx_1) 138, an additional transmit antenna (Tx_2) 140, a firstreceive antenna (Rx_1) 106, and an additional receive antenna (Rx_2)108. The wireless communication system 100 may further comprise a mixer110, an adder 112, an RF block 114, a baseband (BB) processor 119, asingle weight baseband generator (SWBBG) 121, a single weight generator(SWG) channel estimator 122, and a SWG algorithm block 124.

The first transmit antenna, Tx_1 138, and the additional or secondtransmit antenna, Tx_2 140, may comprise suitable hardware that may beadapted to transmit a plurality of SC communication signals, s_(T), froma wireless transmitter device. The first receive antenna, Rx_1 106, andthe additional or second receive antenna, Rx_2 108, may comprisesuitable hardware that may be adapted to receive at least a portion ofthe transmitted SC communication signals in a wireless receiver device.For example, the receive antenna Rx_1 106 may receive signal s_(R1)while the receive antenna Rx_2 108 may receive signal s_(R2). Thepropagation channels that corresponds to the paths taken by the SCcommunication signals transmitted from the transmit antennas Tx_1 138and Tx_2 140 and received by the receive antenna Rx_1 106 may berepresented by h₁₁ and h₁₂ respectively. In this regard, h₁₁ and h₁₂ mayrepresent the actual time varying impulse responses of the radiofrequency (RF) paths taken by the SC communication signals transmittedfrom the transmit antennas Tx_1 138 and Tx_2 140 and received by thereceive antenna Rx_1 106. The actual time varying impulse responses,h_(xy), may contain multiple propagation paths arriving at differentdelays.

Similarly, the propagation channels that corresponds to the paths takenby the SC communication signals transmitted from the transmit antennasTx_1 138 and Tx_2 140 and received by the receive antenna Rx_2 108 maybe represented by h₂₁ and h₂₂ respectively. In this regard, h₂₁ and h₂₂may represent the actual time varying impulse responses of the RF pathstaken by the SC communication signals transmitted from the transmitantennas Tx_1 138 and Tx_2 140 and received by the receive antenna Rx_2108. In some instances, a wireless transmitter device may be adapted toperiodically transmit calibration and/or pilot signal that may beutilized by a 2-Rx antennas wireless receiver device to determineestimates of h₁₁, h₁₂, h₂₁, and h₂₂. The 2-Tx and 2-Rx antennas wirelesscommunication system 100 in FIG. 1A may represent a MIMO communicationsystem.

The mixer 110 may comprise suitable logic and/or circuitry that may beadapted to operate as a complex multiplier that may modify the amplitudeand/or phase of the portion of the SC communication signals received bythe receive antenna Rx_2 108 via a rotation waveform e^(jw) ^(r) _(t)provided by the SWBBG 121, where w_(r)=2πf_(r) and f_(r) is the rotationfrequency. In this regard, a channel weight comprising an amplitudecomponent and phase component may be provided by the SWBBG 121 formodifying the signal received by the receive antenna Rx_2 108 to achievechannel orthogonality between the receive antenna Rx_1 106 and thereceive antenna Rx_2 108. In some implementations, the mixer 110 maycomprise a variable gain amplifier and a phase shifter, for example.

Through the achieved channel orthogonality, estimates of h₁₁, h₁₂, h₂₁,and h₂₂ may be determined by the SWG channel estimator 122 in the SWBBG121. The h₁₁, h₁₂, h₂₁, and h₂₂ estimates may be utilized by the SWGalgorithm block 124 to determine an optimum amplitude A and phase φ thatmodify signals received by the receive antenna Rx_2 108 via mixer 110 sothat the receiver signal-to-interference-and-noise ratio (SINR) ismaximized, for example. In some instances, instead of utilizing therotation waveform e^(jw) ^(r) _(t) to achieve the channel orthogonalitybetween the receive antenna Rx_1 106 and the receive antenna Rx_2 108,square or triangular waveforms may be also utilized. Moreover, waveformsrepresenting different orthogonal codes may also be utilized, similar tothe CDMA orthogonal codes with the same spreading.

The output of the mixer 110 may be transferred to a bandpass filter, alow noise amplifier (LNA), and/or a phase shifter for further processingof the received signals. The adder 112 may comprise suitable hardware,logic, and/or circuitry that may be adapted to add the output of thereceive antenna Rx_1 106 and the output of the mixer 110 to generate acombined received SC communication signal, s_(RC). In some instances,bringing the output signals of the receive antenna Rx_1 106 and themixer 110 together into a single electrical connection may provide thefunctionality of the adder 112. Notwithstanding, an output of the adder112 may be transferred to the RF block 114 for further processing of thecombined received SC communication signal, s_(RC).

The RF block 114 may comprise suitable logic and/or circuitry that maybe adapted to process the combined received SC communication signal,s_(RC). The RF block 114 may perform, for example, filtering,amplification, and/or analog-to-digital (A/D) conversion operations. TheBB processor 119 may comprise suitable logic, circuitry, and/or codethat may be adapted to determine a first baseband combined channelestimate, ĥ₁, which may comprise information regarding propagationchannels h₁₁ and h₂₁. The BB processor 119 may also be adapted toprocess the output of the RF block 114 to determine a second basebandcombined channel estimate, ĥ₂, which may comprise information regardingpropagation channels h₁₂ and h₂₂. The BB processor 119 may also beadapted to determine an estimate of the transmitted SC communicationsignals, ŝ_(T).

The SWBBG 121 may comprise suitable logic, circuitry, and/or code thatmay be adapted to receive the first and second baseband combined channelestimates, ĥ₁ and ĥ₂, from the BB processor 119 and generate phase andamplitude components of the rotation waveform to be applied by the mixer110 to modify the portion of the SC communication signals received bythe receive antenna Rx_2 108, s_(R2). The SWG channel estimator 122 maycomprise suitable logic, circuitry, and/or code that may be adapted toprocess the first and second baseband combined channel estimates, ĥ₁ andĥ₂, generated by the BB processor 119 and may determine a matrix Ĥ_(2×2)of propagation channel estimates ĥ₁₁, ĥ₁₂, ĥ₂₁, and ĥ₂₂, whichcorrespond to estimates of a matrix H_(2×2) of time varying impulseresponses h₁₁, h₁₂, h₂₁, and h₂₂ respectively. The actual time varyingimpulse responses, h_(xy), may contain multiple propagation pathsarriving at different delays. In that regard, the matrix Ĥ_(2×2) ofpropagation channel estimates ĥ₁₁, ĥ₁₂, ĥ₂₁, and ĥ₂₂ may consist ofmultiple path estimates arriving at different delays. The SWG algorithmblock 124 may comprise suitable logic, circuitry, and/or code that maybe adapted to determine a channel weight to be transferred to the mixer110 to modify the signal s_(R2) so that the receiver SINR is maximized.The channel weight to be transferred to the mixer 110 may refer to aphase, φ, and amplitude, A, that results in a maximum SINR.

FIG. 1B is a block diagram of an exemplary two-transmit (2-Tx) andtwo-receive (2-Rx) antennas wireless communication withpre-equalization, in accordance with an embodiment of the invention.Referring to FIG. 1B, there is shown a wireless communication system 131that may differ from the wireless communication system 100 in FIG. 1A inthat the wireless communication system 131 further comprises mixers 130and 132 and a pre-equalizer 125.

The mixers 130 and 132 may comprise suitable logic and/or circuitry thatmay be adapted to multiply a signal to be transmitted, s_(T0), withweight factors W₁ and W₂ respectively. For example, the weight factorsW₁ and W₂ may correspond to phase and/or amplitude component feedbackadjustments that may be generated by the pre-equalizer 125. In thisregard, the pre-equalizer 125 may transfer the weight factors orparameters that correspond to those weight factors to the transmittervia an uplink feedback process.

The pre-equalizer 125 may comprise suitable logic, circuitry, and/orcode that may be adapted to determine a plurality of pre-equalizationparameters and/or weight factors W₁ and W₂ based on the matrix Ĥ_(2×2)of propagation channel estimates ĥ₁₁, ĥ₁₂, ĥ₂₁, and ĥ₂₂. Thepre-equalization parameters may comprise phase and amplitude informationto be transferred to the transmitter portion of the wirelesscommunication system 131. The weights or weight parameters determined bythe pre-equalizer 125 may be a single channel tap or a weight vector fora frequency selective propagation channel. Moreover, the pre-equalizer125 may be adapted to determine the pre-equalization parameters basedon, for example, a least-mean squares (LMS) algorithm, a recursive leastsquares (RLS) algorithm, direct matrix inversion, a cost functionanalysis, or a second order statistical technique. When utilizing a costfunction analysis, for example, coefficients utilized by thepre-equalizer to determine the pre-equalization parameters may beobtained based on the minimization of a cost function, J, of the formJ=f(SINR) or J=f(SNR), where f(x) denotes a function of variable x andSINR and SNR are the signal-to-interference-and-noise ratio andsignal-to-noise ratio of the received signals respectively. For example,a cost function J=(SINR)⁻¹ may be minimized to obtain pre-equalizercoefficients that may be utilized to determine the pre-equalizationparameters. The pre-equalizer may apply and/or modify cost functionparameters associated with variables utilized with the cost function. Incertain instances, pre-coding techniques may be utilized in order torequire less complicated processing of the pre-equalization parameterson the receiver side.

The SWG algorithm block 124 in FIG. 1B may be adapted to supporttwo-transmit antenna closed loop mode 1 (CL1) and closed loop mode 2(CL2) for transmit diversity as described in the 3^(rd) GenerationProject Partnership (3GPP), Technical Specification Group Radio AccessNetwork, Physical Layer Procedures (FDD), Release 6 (3GPP TS 25.214V5.5.0, 2003-06). When either CL1 or CL2 are active, the wirelesscommunication system 131 may be said to be in an active closed loop modeof operation. The SWG algorithm block 124 may generate weight factors W₁and W₂ to support two-transmit antenna CL1 and CL2 transmit diversity.In this regard, the SWG algorithm block 124 may utilize, for example,similar operations as those for determining phase and amplitudeadjustments at the wireless receiver when determining the phase andamplitude adjustments to be applied at a diversity transmitter wheneither CL1 or CL2 are active.

FIG. 1C is a block diagram of an exemplary two-transmit (2-Tx) andtwo-receive (2-Rx) antennas wireless communication system that supportsWCDMA/HSPDA, in accordance with an embodiment of the invention.Referring to FIG. 1C, there is shown a wireless communication system 135that may differ from the wireless communication system 100 in FIG. 1A inthat the wireless communication system 135 may comprise chip matchingfilter (CMF) 116, a cluster path processor (CPP) 118, and a baseband(BB) processor 120.

The CMF 116 may comprise suitable logic, circuitry, and/or code that maybe adapted to operate as a matched-filter on the digital output from theRF block 114. The output of the CMF 116 may be transferred, for example,to the CPP 118 and/or to the BB processor 120 for further processing.The CPP 118 may comprise suitable logic, circuitry, and/or code that maybe adapted to process the filtered output of the CMF 116 to determine afirst baseband combined channel estimate, ĥ₁, which may compriseinformation regarding propagation channels h₁₁ and h₂₁. The CPP 118 mayalso be adapted to process the filtered output of the CMF 116 todetermine a second baseband combined channel estimate, ĥ₂, which maycomprise information regarding propagation channels h₁₂ and h₂₂. In thisregard, the CPP 118 may process the received signals in clusters. U.S.application Ser. No. 11/173,854 provides a detailed description ofsignal clusters and is hereby incorporated herein by reference in itsentirety. The CPP 118 may also be adapted to generate a lock indicatorsignal that may be utilized by, for example, the BB processor 120 as anindication of whether the channel estimates are valid. The BB processor120 may comprise suitable logic, circuitry, and/or code that may beadapted to digitally process the filtered output of the CMF 116 todetermine an estimate of the transmitted SC communication signals,ŝ_(T).

FIG. 1D is a block diagram of an exemplary two-transmit (2-Tx) andtwo-receive (2-Rx) antennas wireless communication system withpre-equalization that supports WCDMA/HSPDA, in accordance with anembodiment of the invention. Referring to FIG. 1D, there is shown awireless communication system 137 that may differ from the wirelesscommunication system 135 in FIG. 1C in that the wireless communicationsystem 137 may comprise a dedicated physical channel (DPCH) block 126, amixer 128, a first combiner 134, a second combiner 136, and thepre-equalizer 125. The pre-equalizer 125 and the SWG algorithm block 124in FIG. 1D may be adapted to operate substantially as the pre-equalizer125 and the SWG algorithm block 124 in FIG. 1B.

The DPCH block 126 may comprise suitable logic, circuitry, and/or codethat may be adapted to receive a plurality of input channels, forexample, a dedicated physical control channel (DPCCH) and a dedicatedphysical data channel (DPDCH). The DPCH 126 may be adapted tosimultaneously control the power on each of the DPCCH and DPDCHchannels. The mixer 128 may comprise suitable logic and/or circuitrythat may be adapted to multiply the output of DPCH 126 with a spreadand/or scramble signal to generate a spread complex-valued signal thatmay be transferred to the inputs of the mixers 130 and 132.

The output of the mixer 130 may be transferred to the first combiner 134and the output of the mixer 132 may be transferred to the secondcombiner 236. The first and second combiners 134 and 136 may comprisesuitable logic, circuitry, and/or code that may be adapted to add orcombine the outputs generated by mixers 130 and 132 with a common pilotchannel 1 (CPICH1) signal and a common pilot channel 2 (CPICH2) signalrespectively. The CPICH1 signal and CPICH2 signals may comprise fixedchannelization code allocation and may be utilized to measure the signalphase and amplitude and strength of the propagation channels between thetransmit antennas and the receive antennas.

FIG. 1E is a block diagram of an exemplary two-transmit (2-Tx) andmultiple-receive (M-Rx) antennas wireless communication system withpre-equalization that supports WCDMA/HSPDA, in accordance with anembodiment of the invention. Referring to FIG. 1E, the wirelesscommunication system 150 may differ from the wireless communicationsystem 137 in FIG. 1D in that (M-1) additional receive antennas (Rx_2108 to Rx_M 109) and (M-1) mixers 110 to 111 may be utilized.

The first transmit antenna, Tx_1 138, and the additional or secondtransmit antenna, Tx_2 140, may comprise suitable hardware that may beadapted to transmit a plurality of SC communication signals, s_(T), froma wireless transmitter device. The propagation channels that correspondto the paths taken by the SC communication signals transmitted from thetransmit antennas Tx_1 138 and Tx_2 140 and received by the receiveantennas Rx_1 106 to Rx_M 109 may be represented by an M×2 matrix,H_(M×2). The matrix H_(M×2) may comprise propagation channels h₁₁ toh_(M1), and h₁₂ to h_(M2). In this regard, h₁₁ to h_(M1) may representthe time varying impulse responses of the RF paths taken by the portionof the transmitted SC communication signals transmitted by transmitantenna Tx_1 138 and received by the receive antennas Rx_1 106 to Rx_M109 respectively. Similarly, h₁₂ to h_(M2) may represent the timevarying impulse responses of the RF paths taken by the portion of thetransmitted SC communication signals transmitted by transmit antennaTx_2 140 and received by the receive antennas Rx_1 106 to Rx_M 109respectively. In some instances, a wireless transmitter devicecomprising a first and a second transmit antenna may be adapted toperiodically transmit calibration and/or pilot signals that may beutilized by an M-Rx antenna wireless receiver device to determineestimates of h₁₁ to h_(M1) and h₁₂ to h_(M2). The 2-Tx and M-Rx antennaswireless communication system 150 in FIG. 1B may represent a MIMOcommunication system.

The CPP 118 in FIG. 1E may be adapted to determine a first basebandcombined channel estimate, ĥ₁, which may comprise information regardingpropagation channels h₁₁ to h_(M1). For example, a portion of ĥ₁ maycomprise information regarding the propagation channels between thetransmit antenna Tx_1 138 and the receive antennas Rx_1 106 and Rx_2108, that is, h₁₁ and h₂₁, while another portion of ĥ₁ may compriseinformation regarding the propagation channels between the transmitantenna Tx_1 138 and the receive antennas Rx_1 106 and Rx_M 109, thatis, h₁₁ and h_(M1).

The CPP 118 in FIG. 1E may also be adapted to determine a secondbaseband combined channel estimate, ĥ₂, which may comprise informationregarding propagation channels h₁₂ to h_(M2). For example, a portion ofĥ₂ may comprise information regarding the propagation channels betweenthe transmit antenna Tx_2 140 and the receive antennas Rx_1 106 and Rx_2108, that is, h₁₂ and h₂₂, while another portion of ĥ₂ may compriseinformation regarding the propagation channels between the transmitantenna Tx_2 140 and the receive antennas Rx_1 106 and Rx_M 109, thatis, h₁₂ and h_(M2). The combined channel estimates ĥ₁ and ĥ₂ may bedetermined, that is, may be separated, in the CPP 118 by utilizing theorthogonal relationship between the common pilot signals CPICH1 andCPICH2 transmitted by the antennas Tx_1 138 and Tx_2 140, respectively.

The SWG channel estimator 122 in FIG. 1E may be adapted to process thefirst and second baseband combined channel estimates, ĥ₁ and ĥ₂,determined by the CPP 118 and may determine a matrix Ĥ_(M×2) ofpropagation channel estimates ĥ₁₁ to ĥ_(M1) and ĥ₁₂ to ĥ_(M2), whichcorrespond to estimates of the matrix H_(M×2) of time varying impulseresponses h₁₁ to h_(M1) and h₁₂ to h_(M2), respectively. The SWGalgorithm block 124 may utilize the contents of the matrix Ĥ_(M×2) todetermine (M-1) channel weights utilized by the mixers 110 to 111 tomodify the portions of the transmitted SC communication signals receivedby the additional receive antennas Rx_2 108 to Rx_M 109 so that thereceiver SINR is maximized, for example. The (M-1) channel weights maycomprise amplitude and phase components, A₁ to A_(M−1) and φ₁ toφ_(M−1), for example, that result in a maximum receiver SINR. Thepre-equalizer 125 in FIG. 1B may be adapted to determine a plurality ofpre-equalization parameters based on the matrix Ĥ_(M×2) of propagationchannel estimates ĥ₁₁ to ĥ_(M1) and ĥ₁₂ to ĥ_(M2).

FIG. 2A is a flow diagram illustrating exemplary steps for channelestimation in a 2-Tx and M-Rx antennas wireless communication system, inaccordance with an embodiment of the invention. Referring to FIG. 2A,after start step 202, in step 204, the SC communication signals, s_(T),may be transmitted from the transmit antennas Tx_1 138 and Tx_2 140 inFIG. 1E. In step 206, the first and additional receive antennas, Rx_1106 to Rx_M 109, may receive a portion of the transmitted SCcommunication signals. In step 208, the signals received by theadditional receive antennas Rx_1 106 to Rx_M 109 may be multiplied by,for example, rotation waveforms, such as sine, square, or triangularwaveforms for example, in the mixers 110 to 111. In this regard, therotation waveforms may have a given set of phase component values. Instep 210, the output of the receive antenna Rx_1 106 and the output ofthe mixers 110 to 111 associated with the additional receive antennasRx_2 108 to Rx_M 109 may be added or combined into the received SCcommunication signal, s_(RC). The combination may occur in the adder112, for example.

In step 212, the CPP 118 may determine the first and second basebandcombined channel estimates, ĥ₁ and ĥ₂, which comprise informationregarding propagation channels h₁₁ to h_(M1) and h₁₂ to h_(M2). In step214, the SWG channel estimator 122 in the SWBBG 121 may determine thematrix Ĥ_(M×2) of propagation channel estimates ĥ₁₁ to ĥ_(M1) and ĥ₁₂ toĥ_(M2). In this regard, the propagation channel estimates ĥ₁₁ to ĥ_(M1)and ĥ₁₂ to ĥ_(M2) may be determined concurrently. In step 216, thepre-equalizer 125 may calculate or determine the pre-equalization weightparameters or weight factors W₁ and W₂ that may be applied to the mixers130 and 132 in FIG. 1E respectively. The pre-equalization weights W₁ andW₂ may be transferred to a transmitter, such as a base station, topre-equalize the signals being transmitted from the transmit antennasTx_1 138 and Tx_2 140.

In step 218, the wireless communication system 150 may determine whethera closed loop operating mode that supports transmit diversity modes CL1and CL2 is active. When the closed loop operating mode is active, theprocess may proceed to step 224. In step 224, the (M-1) maximum SINRchannel weights that comprise amplitude and phase components, A₁ toA_(M−1) and φ₁ to φ_(M−1), may be generated concurrently with thediversity weight parameters supported by CL1 or CL2. In this regard, theSWG algorithm block 124 may be utilized to generate the amplitude andphase components and the diversity weight parameters W₁ and W₂. Thechannel weights may be based on the propagation channel estimatesdetermined after the application of pre-equalization weight parametersW₁ and W₂ to the transmitter. The diversity weight parameters supportedby CL1 or CL2 may be transferred to a transmitter, such as a basestation, to combine the signals being transmitted from the transmitantennas Tx_1 138 and Tx_2 140. After step 224, the process may proceedto step 222.

Returning to step 218, when the closed loop operating mode is notactive, the process may proceed to step 220. In step 220, the SWGalgorithm block 124 may generate the (M-1) maximum SINR channel weightsthat comprise amplitude and phase components, A₁ to A_(M−1) and φ₁ toφ_(M−1). In step 222, the (M-1) maximum SINR channel weights may beapplied to the mixers 110 to 111 in FIG. 1E.

After steps 222 or 224, the process may proceed to end step 226 whereadditional SC communication signals received may be phase and amplitudeadjusted based on the maximum SINR channel weights applied to the mixers110 to 111. The channel estimation phase rotation and the maximum SINRphase/amplitude adjustment described in flow chart 200 may be performedcontinuously or may be performed periodically. In this regard, FIG. 2Billustrates an exemplary periodic phase rotation for an in-phase (I)signal received in one of the additional receive antennas, in accordancewith an embodiment of the invention.

FIG. 3A is a block diagram of an exemplary single weight basebandgenerator (SWBBG) that may be utilized in a 2-Tx and 2-Rx antennassystem, in accordance with an embodiment of the invention. Referring toFIG. 3A, a receiver system 300 may correspond to a portion of thewireless communication system 137 in FIG. 1D and may comprise a firstreceive antenna (Rx_1) 302, an additional receive antenna (Rx_2) 304, anadder 306, a mixer 308, a single weight baseband generator (SWBBG) 310,and a pre-equalizer 322. The SWBBG 310 may comprise a phase rotatorstart controller 314, a delay block 316, a single weight generator (SWG)channel estimator 318, an SWG algorithm block 320, and an RF phase andamplitude controller 312. The SWBBG 310 may represent an exemplaryimplementation of the SWBBG 121 in FIG. 1D.

The first receive antenna, Rx_1 302, and the additional or secondreceive antenna, Rx_2 304, may comprise suitable hardware that may beadapted to receive at least a portion of transmitted SC communicationsignals in the receiver system 300. For example, the receive antennaRx_1 302 may receive a signal s_(R1) while the receive antenna Rx_2 304may receive a signal s_(R2). The mixer 308 may correspond to, forexample, the mixer 110 in FIG. 1D. In some instances, the output of themixer 308 may be communicated to a bandpass filter and/or a low noiseamplifier (LNA) for further processing of the received signals.

The adder 306 may comprise suitable hardware, logic, and/or circuitrythat may be adapted to add the output of the receive antenna Rx_1 302and the output of the mixer 308 to generate a combined received SCcommunication signal, s_(RC). In some instances, bringing the outputsignals of the receive antenna Rx_1 302 and the mixer 308 together intoa single electrical connection may provide the functionality of theadder 306. The output of the adder 306 may be transferred to additionalprocessing blocks for RF and baseband processing of the combinedreceived SC communication signal, s_(RC).

The phase rotator and start controller 314 may comprise suitable logic,circuitry, and/or code that may be adapted to control portions of theoperation of the RF phase and amplitude controller 312 and to controlthe delay block 316. The phase rotator and start controller 314 mayreceive a signal, such as a reset signal, from, for example, the BBprocessor 120 in FIG. 1D, or from firmware operating in a processor, toindicate the start of operations that determine the propagation channelestimates and/or the channel weight to apply to the mixer 308. The delayblock 316 may comprise suitable logic, circuitry, and/or code that maybe adapted to provide a time delay to compensate for the RF/modem delay.The delay may be applied in order to compensate for the interval of timethat may occur between receiving the combined channel estimates, ĥ₁ andĥ₂, modified by the rotation waveform and the actual rotating waveformat the mixer 308.

The SWG channel estimator 318 may comprise suitable logic, circuitry,and/or code that may be adapted to process the first and second basebandcombined channel estimates, ĥ₁ and ĥ₂, and determine the matrix Ĥ_(2×2)of propagation channel estimates ĥ₁₁, ĥ₁₂, ĥ₂₁, and ĥ₂₂. The SWG channelestimator 318 may also be adapted to generate an algorithm start signalto the SWG algorithm block 320 to indicate that the propagation channelestimates ĥ₁₁, ĥ₁₂, ĥ₂₁, and ĥ₂₂ are available for processing. In thisregard, the algorithm start signal may be asserted when integrationoperations performed by the SWG channel estimator 318 have completed.

The SWG algorithm block 320 may comprise suitable logic, circuitry,and/or code that may be adapted to determine a channel weight to betransferred to the mixer 308 via the RF phase and amplitude controller312 to modify the signal s_(R2). The channel weight to be transferred tothe mixer 308 may refer to the phase, φ, and amplitude, A. The channelweight may be based on the propagation channel estimates ĥ₁₁, ĥ₁₂, ĥ₂₁,and ĥ₂₂ and on additional information such as noise power estimates andinterference propagation channel estimates, for example. The SWGalgorithm block 320 may also be adapted to generate an algorithm endsignal to indicate to the RF phase and amplitude controller 312 that thechannel weight has been determined and that it may be applied to themixer 308.

The SWG algorithm block 320 may also be adapted to generate a portion ofthe weight parameters or weight factors W₁ and W₂ related to the closedloop diversity operation. The channel weights and closed loop diversityweights may be calculated jointly to maximize the receiver SINR, forexample. The pre-equalizer 322 may comprise suitable logic, circuitry,and/or code that may be adapted to determine a plurality ofpre-equalization parameters based on the matrix Ĥ_(2×2) of propagationchannel estimates ĥ₁₁, ĥ₁₂, ĥ₂₁, and ĥ₂₂. The pre-equalizer 322 may alsobe adapted to generate a portion of the weight parameters or weightfactors W₁ and W₂. In this regard, the pre-equalizer 322 may generatethe weight factors W₁ and W₂ when the closed loop operating mode is notactive, while the SWG algorithm block 320 may generated the weightfactors W₁ and W₂ when the closed loop operating mode is active, forexample.

The RF phase and amplitude controller 312 may comprise suitable logic,circuitry, and/or code that may be adapted to apply the rotationwaveform e^(jw) ^(r) _(t) to the mixer 308. When phase and amplitudecomponents, A and φ, that correspond to the channel weight determined bythe SWG algorithm block 320 are available, the RF phase and amplitudecontroller 312 may apply amplitude A and phase φ to the mixer 308. Inthis regard, the RF phase and amplitude controller 312 may apply therotation waveform or the amplitude and phase components in accordancewith the control signals provided by the phase rotator start controller314 and/or the algorithm end signal generated by the SWG algorithm block320.

The phase rotation operation performed on the s_(R2) signal in theadditional receive antenna Rx_2 304 may be continuous or periodic. Acontinuous rotation of the s_(R2) signal may be perceived by a wirelessmodem as a high Doppler, and for some modem implementations this maydecrease the modem's performance. When a periodic rotation operation isutilized instead, the period between consecutive phase rotations maydepend on the Doppler frequency perceived by the wireless modem. Forexample, in a higher Doppler operation, it may be necessary to performmore frequent channel estimation while in a lower Doppler operation,channel estimation may be less frequent. The signal rotation period mayalso depend on the desired wireless modem performance and the accuracyof the propagation channel estimation. For example, when the Dopplerfrequency is 5 Hz, the period between consecutive rotations may be 1/50sec., that is, 10 rotations or channel estimations per signal fade.

FIG. 3B is a block diagram of an exemplary single weight basebandgenerator (SWBBG) that may be utilized in a 2-Tx and M-Rx antennassystem, in accordance with an embodiment of the invention. Referring toFIG. 3B, a receiver system 330 may correspond to a portion of thewireless communication system 150 in FIG. 1E and may differ from thereceiver system 300 in FIG. 3A in that (M-1) additional receiveantennas, Rx_2 304 to Rx_M 305, and (M-1) mixers 308 to 309 may beutilized. In this regard, the SWG channel estimator 318 may be adaptedto process the first and second baseband combined channel estimates, ĥ₁and ĥ₂, and determine the matrix Ĥ_(M×2) of propagation channelestimates ĥ₁₁ to ĥ_(M1) and ĥ₁₂ to ĥ_(M2).

The SWG algorithm block 320 may also be adapted to determine (M-1)channel weights, that may be utilized to maximize receiver SINR, forexample, to be applied to the mixers 308 to 309 to modify the portionsof the transmitted SC communication signals received by the additionalreceive antennas Rx_2 304 to Rx_M 305. The (M-1) channel weights maycomprise amplitude and phase components, A₁ to A_(M−1) and φ₁ toφ_(M−1). The RF phase and amplitude controller 312 may also be adaptedto apply rotation waveforms e^(jw) ^(r1) ^(t) to e^(jw) ^(r(M−1)) ^(t)or phase and amplitude components, A₁ to A_(M−1) and φ₁ to φ_(M−1), tothe mixers 308 to 309. In this regard, the RF phase and amplitudecontroller 312 may apply the rotation waveforms or the amplitude andphase components in accordance with the control signals provided by thephase rotator start controller 314 and/or the algorithm end signalgenerated by the SWG algorithm block 320. The SWG algorithm block 320may also be adapted to generate a portion of the weight parameters orweight factors W₁ and W₂ related to the closed loop diversity operation.The channel weights and closed loop diversity weights may be calculatedjointly to maximize the receiver SINR, for example. The pre-equalizer322 in FIG. 3B may also be adapted to determine a plurality ofpre-equalization parameters based on the matrix Ĥ_(M×2) of propagationchannel estimates ĥ₁, to ĥ_(M1) and ĥ₁₂ to ĥ_(M2).

FIG. 3C is a block diagram of an exemplary RF phase and amplitudecontroller, in accordance with an embodiment of the invention. Referringto FIG. 3C, the RF phase and amplitude controller 312 may comprise aswitch 340, a plurality of rotation waveform sources 342, and aplurality of SWG algorithm weights 344. The switch 340 may comprisesuitable hardware, logic, and/or circuitry that may be adapted to selectbetween the rotation waveforms e^(jw) ^(r1) ^(t) to e^(jw) ^(r(M−1))^(t) and the SWG algorithm determined weights A₁e^(jφ) ¹ toA_(M−1)e^(jφ) ^(M−1) . The rotation waveform sources 342 may comprisesuitable hardware, logic and/or circuitry that may be adapted togenerate the signal e^(jw) ^(rk) ^(t), where w_(rk)=2πf_(rk) and f_(rk)is the rotation frequency that preserves the orthogonality of thereceived signals at the receive antennas Rx_2 302 to Rx_M 305 in FIG.3B, for example. The rotation frequency that preserves the signalorthogonality at the receiving antennas may be selected as w_(rk)=kw_(r)where k=1, 2, . . . , M-1. Other rotation waveforms such as triangularor square waveforms may be utilized with the same frequencyrelationships. Moreover, waveforms representing different orthogonalcodes of the same frequency may also be utilized, similar to the CDMAorthogonal codes with the same spreading. In this embodiment, the signale^(jw) ^(rk) ^(t) may be utilized as an exemplary waveform. Theplurality of SWG algorithm weights 344 may comprise suitable hardware,logic, and/or circuitry that may be adapted to generate the signalsA₁e^(jφ) ¹ to A_(M−1)e^(jφ) ^(M−1) from the amplitude and phasecomponents, A₁ to A_(M−1) and φ₁ to φ_(M−1), respectively.

In operation, the RF phase and amplitude controller 312 may apply thesignals e^(jw) ^(r1) ^(t) to e^(jw) ^(r(M−1)) ^(t) to the mixers 308 to309 in FIG. 3B based on control information provided by the phaserotator start controller 314. The switch 340 may select the rotationwaveform sources 342 based on the control information provided by thephase rotator start controller 314. Once the channel weights aredetermined by the SWG algorithm block 320 and the phase and amplitudecomponents have been transferred to the RF phase and amplitudecontroller 312, the algorithm end signal may be utilized to change theselection of the switch 340. In this regard, the switch 340 may beutilized to select and apply the signals A₁e^(jφ) ¹ to A_(M-1)e^(jφ)^(M−1) to the mixers 308 to 309 in FIG. 3B.

FIG. 4 is a flow diagram illustrating exemplary steps in the operationof the single weight baseband generator (SWBBG) that may be utilized ina 2-Tx and M-Rx antennas system, in accordance with an embodiment of theinvention. Referring to FIG. 4, after start step 402, in step 404, thephase rotator start controller 314 in FIG. 3B may receive the resetsignal to initiate operations for determining propagation channelestimates and channel weights in the SWBBG 310. The phase rotator startcontroller 314 may generate control signals to the delay block 316 andto the RF phase and amplitude controller 312. The control signals to thedelay block 316 may be utilized to determine a delay time to be appliedby the delay block 316. The control signals to the RF phase andamplitude controller 312 may be utilized to determine when to apply therotation waveforms or the channel weights determined by the SWGalgorithm block 124 to the mixers 308 to 309 in FIG. 3B, for example.

In step 406, the RF phase and amplitude controller 312 may apply thesignals e^(jw) ^(r1) ^(t) to e^(jw) ^(r(M−1)) ^(t) to the mixers 308 to309 in FIG. 3B. In step 408, the delay block 316 may apply a time delaysignal to the SWG channel estimator 318 to reflect the interval of timethat may occur between receiving the combined channel estimates, ĥ₁ andĥ₂, modified by the rotation waveform and the actual rotating waveformat the mixer 308. For example, the time delay signal may be utilized asan enable signal to the SWG channel estimator 318, where the assertionof the time delay signal initiates operations for determiningpropagation channel estimates. In step 410, the SWG channel estimator318 may process the first and second baseband combined channelestimates, ĥ₁ and ĥ₂, and may determine the matrix Ĥ_(M×2) ofpropagation channel estimates ĥ₁₁ to ĥ_(M1) and ĥ₁₂ to ĥ_(M2). The SWGchannel estimator 318 may transfer the propagation channel estimates ĥ₁₁to ĥ_(M1) and ĥ₁₂ to ĥ_(M2) to the SWG algorithm block 320. In step 412,the pre-equalizer 322 may calculate or generate the pre-equalizationweight parameters or weight factors W₁ and W₂. The pre-equalizationweight parameters may be transferred to a wireless transmitter, such asa base station.

In step 414, the receiver system 330 in FIG. 3B may determine whether aclosed loop operating mode that supports transmit diversity modes CL1and CL2 is active. When the closed loop operating mode is active, theprocess may proceed to step 418. In step 418, the (M-1) maximum SINRchannel weights that comprise amplitude and phase components, A₁ toA_(M−1) and φ₁ to φ_(M−1), may be generated by the SWG algorithm block320 concurrently with the diversity weight parameters W₁ and W₂supported by CL1 or CL2. The channel weights may be based on thepropagation channel estimates determined after the application ofpre-equalization weight parameters W₁ and W₂ to the transmitter. Thediversity weight parameters that support CL1 or CL2 may be transferredto a transmitter, such as a base station, to apply the weights to thesignals being transmitted. After step 418, the process may proceed tostep 420.

Returning to step 414, when the closed loop operating mode is notactive, the process may proceed to step 416. In step 416, the SWGalgorithm block 320 may generate the (M-1) maximum SINR channel weightsthat comprise amplitude and phase components, A₁ to A_(M−1) and φ₁ toφ_(M−1), based on the propagation channel estimates ĥ₁₁ to ĥ_(M1) andĥ₁₂ to ĥ_(M2) and/or noise power estimates and interference channelestimates, for example. The SWG algorithm block 320 may transfer thechannel weights to the RF phase and amplitude controller 312. The SWGalgorithm block 320 may generate the algorithm end signal to indicate tothe RF phase and amplitude controller 312 that the channel weights areavailable to be applied to the mixers 308 to 309. In step 420, RF phaseand amplitude controller 312 may apply the maximum SINR weights withphase and amplitude components, A₁ to A_(M−1) and φ₁ to φ_(M−1), to themixers 308 to 309 in FIG. 3B, in accordance with the control signalsprovided by the phase rotator start controller 314 and/or the SWGalgorithm block 320.

In step 422, the receiver system 330 in FIG. 3B may determine whetherthe phase rotation operation on the received SC communication signals isperiodic. When the phase rotation operation is not periodic butcontinuous, the process may proceed to step 408 where a new delay may beapplied to the SWG channel estimator 318. In instances when the phaserotation operation is periodic, the process may proceed to step 424where the receiver system 330 may wait until the next phase rotationoperation is initiated by the reset signal. In this regard, the processmay return to step 404 upon assertion of the reset signal on the phaserotator start controller 314.

FIG. 5 is a flow diagram illustrating exemplary steps for determiningchannel weights in additional receive antennas utilizing signal-to-noiseratio (SNR) or signal-to-interference-and-noise ratio (SINR), inaccordance with an embodiment of the invention. Referring to FIG. 5,after start step 502, in step 504, the SWG algorithm block 320 maydetermine whether the signals received in the receive antennas are noiselimited. The SWG algorithm block 320 may receive noise statistics and/orother noise information from either the CPP 118 and/or from the BBprocessor 120. When the received signals are noise limited, the flowdiagram control may proceed to step 508. In step 508, the SWG algorithmblock 320 may generate models for the received signals. For example, themodels for a 1-Tx and 2-Rx antennas system may be represented by thefollowing expressions:r ₁ =h ₁ s+n ₁,r ₂ =Ae ^(jθ) h ₂ s+Ae ^(jθ) n ₂, andy=r ₁ +r ₂ =s(h ₁ +Ae ^(jθ) h ₂)+n ₁ +Ae ^(jθ) n ₂,where r₁ may represent a model of the signal received in a first receiveantenna, r₂ may represent a model of the signal received in the secondreceive antenna, s may represent the transmitted signal, and n₁ mayrepresent a noise component at the first receive antenna, whose timevarying impulse response is represented by h₁. The parameter n₂ mayrepresent a noise component at the second receive antenna, whose timevarying impulse response is represented by h₂, θ may represent the phasefactor between the signal received in the first and second receiveantennas, and A may represent an amplitude factor. The parameter y mayrepresent the sum of the received signal models and may comprise acombined signal component s(h₁+Ae^(jθ)h₂) and a combined noise componentn₁+Ae^(jθ)n₂.

In step 510, the received signal models may be utilized to determine asignal strength parameter. In this regard, the signal-to-noise ratio(SNR) may correspond to the signal strength parameter to be determined.For example, for a 1-Tx and 2-Rx antennas system, the SNR may bedetermined by maximizing the following expression for various phase, θ,and amplitude, A, factors:

${SNR} = {\frac{{{h_{1} + {A\;{\mathbb{e}}^{j\;\vartheta}h_{2}}}}^{2}}{{E{n_{1}}^{2}} + {E{{A\;{\mathbb{e}}^{j\;\vartheta}n_{2}}}^{2}}} = {\frac{{{h_{1} + {A\;{\mathbb{e}}^{j\;\vartheta}h_{2}}}}^{2}}{\sigma^{2}\left( {1 + A^{2}} \right)}.}}$The SNR numerator may correspond to the y parameter's combined signalcomponent while the SNR denominator may correspond to the y parameter'scombined noise component. The phase factor, θ, may be selected, forexample, from a 360-degrees phase rotation while the amplitude factor,A, may be selected, for example, from an set amplitude range. In oneembodiment of the invention, the phase factor may be varied in aplurality of phase factor steps over the 360-degrees phase rotation tofind the maximum SNR value. In another embodiment of the invention, thephase factor may be varied in a plurality of phase factors steps overthe 360-degrees phase rotation and the amplitude factor may be varied ina plurality of amplitude factor values over the amplitude range to findthe maximum SNR value.

In step 520, after determining the maximum SNR in step 510, the SWGalgorithm block 320 may utilize the amplitude factor and phase factorthat corresponds to the maximum SNR to determine the amplitude and phaseto be provided to the RF amplitude and phase controller 312 in step 520.For example, in one embodiment of the invention, the amplitude and/orphase factors that correspond to the maximum SNR may be utilized as theamplitude and phase to be transferred to the RF amplitude and phasecontroller 312. After application of the appropriate amplitude and phaseby the RF amplitude and phase controller 312 to the receive antennas,the flow diagram control may proceed to end step 522 until a next phaseand amplitude determination is necessary.

Returning to step 504, when received signals are not noise limited, theflow control may proceed to step 506 where a determination may be madeas to whether multiple interfering signals may be present and may needto be considered during channel weight determination. When a singleinterferer is considered, the flow diagram control may proceed to step512. In step 512 the SWG algorithm block 320 may generate models for thereceived signals. For example, the models for a 1-Tx and 2-Rx antennassystem may be represented by the following expressions:r ₁ =h ₁ s+ _(I1) s _(I) +n ₁,r ₂ =Ae ^(jθ)(h ₂ s+h _(I2) s _(I) +n ₂), andy=r ₁ +r ₂ =s(h ₁ +Ae ^(jθ) h ₂)+n ₁ +s _(I)(h _(I1) +Ae ^(jθ) h_(I2))+Ae ^(jθ) n ₂,where r₁ may represent a model of the signal received in a first receiveantenna, r₂ may represent a model of the signal received in the secondreceive antenna, s may represent the transmitted signal, s_(I) mayrepresent the interference signal, and n₁ may represent a noisecomponent at the first receive antenna whose time varying impulseresponse is h₁. The parameter n₂ may represent a noise component at thesecond receive antenna whose time varying impulse response is h₂, θ mayrepresent the phase factor between the signal received in the first andsecond receive antennas, and A may represent an amplitude factor.Moreover, the time varying impulse response h_(I1), may correspond tothe propagation channel between the interference signal source and thefirst receive antenna and the time varying impulse response h_(I2) maycorrespond to the propagation channel between the interference signalsource and the second receive antenna. The parameter y may represent thesum of the received signal models and may comprise a combined signalcomponent s(h₁+Ae^(jθ)h₂) and a combined noise plus interferencecomponent n₁+s_(I)(h_(I1)+Ae^(jθ)h_(I2))+Ae^(jθ)n₂.

In step 514, the received signal models may be utilized to determine asignal strength parameter. In this regard, thesignal-to-interference-and-noise ratio (SINR) may correspond to thesignal strength parameter to be determined. For example, for a 1-Tx and2-Rx antennas system, the SINR may be determined by maximizing thefollowing expression for various phase, θ, and amplitude, A, factors:

${SINR} = {\frac{{{h_{1} + {A\;{\mathbb{e}}^{j\;\vartheta}h_{2}}}}^{2}}{{E{n_{1}}^{2}} + {E{{A\;{\mathbb{e}}^{j\;\vartheta}n_{2}}}^{2}} + {{h_{I\; 1} + {A\;{\mathbb{e}}^{j\;\vartheta}h_{I\; 2}}}}^{2}}\mspace{56mu} = {\frac{{{h_{1} + {A\;{\mathbb{e}}^{j\;\vartheta}h_{2}}}}^{2}}{{\sigma^{2}\left( {1 + A^{2}} \right)} + {{h_{I\; 1} + {A\;{\mathbb{e}}^{j\;\vartheta}h_{I\; 2}}}}^{2}}.}}$where σ² is the noise power. The above SINR equations may be easilyextended to the SC MIMO case. The transmit antennas may include CL1 orCL2 transmit diversity weights. The joint transmit-received solution maybe formed in that case that may include the transmit CL weights and theadditional transmit antenna channel components in the SINR numerator.The SINR numerator may correspond to the y parameter's combined signalcomponent while the SINR denominator may correspond to the y parameter'scombined noise plus interference component. The phase factor, θ, may beselected, for example, from a 360-degrees phase rotation while theamplitude factor, A, may be selected, for example, from an set amplituderange. In one embodiment of the invention, the phase factor may bevaried in a plurality of phase factor steps over the 360-degrees phaserotation to find the maximum SNR value. In another embodiment of theinvention, the phase factor may be varied in a plurality of phasefactors steps over the 360-degrees phase rotation and the amplitudefactor may be varied in a plurality of amplitude factor values over arange of amplitudes to find the maximum SINR value.

After determining the SINR in step 514, the SWG algorithm block 320 maydetermine the amplitude and phase to be provided to the RF amplitude andphase controller 312 in step 520. After application of the appropriateamplitude and phase by the RF amplitude and phase controller 312, theflow diagram control may proceed to end step 522 until a next phase andamplitude determination is necessary.

After determining the SINR in step 518, the SWG algorithm block 320 maydetermine the amplitude and phase to be provided to the RF amplitude andphase controller 312 in step 520. After application of the appropriateamplitude and phase by the RF amplitude and phase controller 312, theflow diagram control may proceed to end step 522 until a next phase andamplitude determination is necessary.

The operations to maximize the signal strength described for steps 510,514, and 518 may be based on a search algorithm. In an exemplaryembodiment of the invention, a search algorithm may be utilized tosearch over 360-degrees phase rotation in 45 or 90 degree phase factorsteps and over a 0-5 amplitude range in 0.25 amplitude values or steps,for example. For a 1-Tx and 2-Rx antenna system, with 90-degree phasefactor steps, a phase only search algorithm may calculate 4 SNR or SINRvalues, for example. For a 2-Tx and 2-Rx antenna system with STTDtransmit mode, with 90-degree phase factor steps, a phase only searchalgorithm may calculate 4 SNR or SINR values. For a 2-Tx and 2-Rxantenna system with the CL1 diversity mode, with 90-degree phase factorsteps at both receiver and transmitter, a phase only search algorithmmay calculate 4×4=16 SNR or SINR values. For a 2-Tx and 2-Rx antennasystem with the CL2 diversity mode, with 90-degree phase factor steps atthe receiver and 45-degree phase factor steps and two power scalingweight levels at the transmitter, a phase only search algorithm maycalculate 4×8×2=64 SNR or SINR values, for example. The maximum valuegenerated by the algorithm may be the output of the search algorithm.

In another embodiment of the invention, a closed-form mathematicalexpression may also be utilized to maximize the SNR and/or the SINR.Utilizing an algorithm or closed-form expression that maximizes the SINRor SNR may provide a good compromise between implementation complexityand performance gains. Notwithstanding, the invention is not limited inthis regard, and other channel weight algorithms may also be utilized.

Determining channel weights and/or pre-equalization parameters may beperformed by monitoring the baseband combined channel estimates, forexample. In this regard, a SWBBG may be utilized for monitoring thebaseband combined channel estimates generated by, for example, a CPP.U.S. application Ser. No. 11/174,252 provides a detailed description ofmonitoring baseband combined channel estimates and is herebyincorporated herein by reference in its entirety.

Another embodiment of the invention may provide a machine-readablestorage, having stored thereon, a computer program having at least onecode section executable by a machine, thereby causing the machine toperform the steps as described above for pre-equalization in a singleweight, single channel MIMO system.

The approach described herein for determining pre-equalizationparameters in a single channel MIMO system may provide a good compromisebetween implementation complexity and performance gains in the designand operation of MIMO systems.

Accordingly, the present invention may be realized in hardware,software, or a combination of hardware and software. The presentinvention may be realized in a centralized fashion in at least onecomputer system, or in a distributed fashion where different elementsare spread across several interconnected computer systems. Any kind ofcomputer system or other apparatus adapted for carrying out the methodsdescribed herein is suited. A typical combination of hardware andsoftware may be a general-purpose computer system with a computerprogram that, when being loaded and executed, controls the computersystem such that it carries out the methods described herein.

The present invention may also be embedded in a computer programproduct, which comprises all the features enabling the implementation ofthe methods described herein, and which when loaded in a computer systemis able to carry out these methods. Computer program in the presentcontext means any expression, in any language, code or notation, of aset of instructions intended to cause a system having an informationprocessing capability to perform a particular function either directlyor after either or both of the following: a) conversion to anotherlanguage, code or notation; b) reproduction in a different materialform.

While the present invention has been described with reference to certainembodiments, it will be understood by those skilled in the art thatvarious changes may be made and equivalents may be substituted withoutdeparting from the scope of the present invention. In addition, manymodifications may be made to adapt a particular situation or material tothe teachings of the present invention without departing from its scope.Therefore, it is intended that the present invention not be limited tothe particular embodiment disclosed, but that the present invention willinclude all embodiments falling within the scope of the appended claims.

1. A method for handling wireless communication comprising: performingusing a processor, circuit, or combination thereof: receiving via afirst receive antenna and an additional receive antenna, pre-equalizedsingle channel (SC) current communication signals generated based onpreviously communicated pre-equalization weight parameters; andmodifying pre-equalized SC subsequent communication signals received viathe additional receive antenna utilizing a plurality of channel weightsthat are derived from the received pre-equalized SC currentcommunication signals.
 2. The method according to claim 1, comprisingdetermining a plurality of current channel estimates based on thereceived pre-equalized SC current communication signals.
 3. The methodaccording to claim 2, comprising determining the plurality of channelweights based on the determined plurality of current channel estimates.4. The method according to claim 1, wherein the previously communicatedpre-equalization weight parameters correspond to signals previouslyreceived by at least one of the first receive antenna or the additionalreceive antenna.
 5. The method according to claim 4, comprisingdetermining a plurality of channel estimates based on the signalspreviously received by at least one of the first receive antenna or theadditional receive antenna.
 6. The method according to claim 5,comprising determining the previously communicated pre-equalizationweight parameters based on the determined plurality of channelestimates.
 7. The method according to claim 1, comprising determiningclosed loop diversity weight parameters and the plurality of channelweights concurrently, if a closed loop mode of operation is active. 8.The method according to claim 1, comprising determining the previouslycommunicated pre-equalization weight parameters based on a least meansquares algorithm, a recursive least squares algorithm, or direct matrixinversion.
 9. A non-transitory machine-readable storage medium havingstored thereon, a computer program having a code section for handlingwireless communication, the code section being executable by a machinefor causing the machine to perform steps comprising: receiving via afirst receive antenna and an additional receive antenna, pre-equalizedsingle channel (SC) current communication signals generated based onpreviously communicated pre-equalization weight parameters; andmodifying pre-equalized SC subsequent communication signals received viathe additional receive antenna utilizing a plurality of channel weightsthat are derived from the received pre-equalized SC currentcommunication signals.
 10. The non-transitory machine-readable storagemedium according to claim 9, wherein the code section comprises code fordetermining a plurality of current channel estimates based on thereceived pre-equalized SC current communication signals.
 11. Thenon-transitory machine-readable storage medium according to claim 10,wherein the code section comprises code for determining the plurality ofchannel weights based on the determined plurality of current channelestimates.
 12. The non-transitory machine-readable storage mediumaccording to claim 9, wherein the previously communicatedpre-equalization weight parameters correspond to signals previouslyreceived by at least one of the first receive antenna or the additionalreceive antenna.
 13. The non-transitory machine-readable storage mediumaccording to claim 12, wherein the code section comprises code fordetermining a plurality of channel estimates based on the signalspreviously received by at least one of the first receive antenna or theadditional receive antenna.
 14. The non-transitory machine-readablestorage medium according to claim 13, wherein the code section comprisescode for determining the previously communicated pre-equalization weightparameters based on the determined plurality of channel estimates. 15.The non-transitory machine-readable storage medium according to claim 9,wherein the code section comprises code for determining closed loopdiversity weight parameters and the plurality of channel weightsconcurrently, if a closed loop mode of operation is active.
 16. Thenon-transitory machine-readable storage medium according to claim 9,wherein the code section comprises code for determining said previouslycommunicated pre-equalization weight parameters based on a least meansquares algorithm, a recursive least squares algorithm, or direct matrixinversion.
 17. A system for handling wireless communication comprising:a circuit, processor, or combination thereof configured to receive via afirst receive antenna and an additional receive antenna, pre-equalizedsingle channel (SC) current communication signals generated based onpreviously communicated pre-equalization weight parameters; and thecircuit processor, or combination thereof configured to modifypre-equalized SC subsequent communication signals received via theadditional receive antenna utilizing a plurality of channel weights thatare derived from the received pre-equalized SC current communicationsignals.
 18. The system according to claim 17, wherein the circuit,processor, or combination thereof is configured to determine a pluralityof current channel estimates based on the received pre-equalized SCcurrent communication signals.
 19. The system according to claim 18,wherein the circuit, processor, or combination thereof is configured todetermine the plurality of channel weights based on the determinedplurality of current channel estimates.
 20. The system according toclaim 17, wherein the previously communicated pre-equalization weightparameters correspond to signals previously received by at least one ofthe first receive antenna or the additional receive antenna.
 21. Thesystem according to claim 20, wherein the circuit, processor, orcombination thereof is configured to determine a plurality of channelestimates based on the signals previously received by at least one ofthe first receive antenna or the additional receive antenna.
 22. Thesystem according to claim 21, wherein the circuit, processor, orcombination thereof is configured to determine the previouslycommunicated pre-equalization weight parameters based on the determinedplurality of channel estimates.
 23. The system according to claim 17,wherein the circuit, processor, or combination thereof is configured todetermine closed loop diversity weight parameters and the plurality ofchannel weights concurrently, if a closed loop mode of operation isactive.
 24. The system according to claim 17, wherein the circuit,processor, or combination thereof is configured to determine thepreviously communicated pre-equalization weight parameters based on aleast mean squares algorithm, a recursive least squares algorithm, ordirect matrix inversion.